1. Field of the Invention
The present invention relates to a balanced photoreceiver and more particularly to a balanced photoreceiver which includes one or more photodiodes directly coupled to an amplifier having a common base input stage which can operate over a frequency band from DC to millimeter wave frequencies with superior output waveform symmetry and relatively low noise performance which can be monolithically integrated to form a balanced receiver monolithic microwave integrated circuit (MMIC).
2. Description of the Prior Art
Balanced photoreceivers for use in, for example, high-data rate communication systems, such as satellite communications, are known. Such balanced photoreceivers are known to be configured for differential phase shift keying (DPSK) which provides increased sensitivity over conventional, single-ended on-off keying photoreceiver topologies. An example of such a balanced photoreceiver is illustrated in FIG. 1. As shown in FIG. 1, the photoreceiver, generally identified with the reference numeral 20, includes a pair of photodiodes 22, 24 and a transimpedance amplifier 26, coupled to the photodiodes 22, and 24 by way of an AC coupling capacitor 28. In such a balanced photoreceiver 20, a phase modulated optical signal from an optical transmitter is optically pre-amplified and demodulated using a MachZehnder interferometer which outputs two complementary optical signals, as generally discussed in: "High Sensitivity Optically Pre-Amplified Direct Detection DPSK Receiver With Active Delay Line Stabilization", by E. Swanson, J. Livas and R. Bondurant, Photonics Technical Letter, vol. 6, No. 2, February 1994, pp. 263-265, hereby incorporated by reference. The two complementary output signals from the MachZehnder interferometer are independently detected by the photodiodes 22 and 24, whose polarities are configured such that if one photodiode 22, 24 is optically excited, it will produce a photocurrent in a direction such that a transimpedance amplifiers 26 output voltage will produce a high voltage level, and if the other photodiode 22, 24 is excited, it will produce a photocurrent in the opposite direction such that the amplifiers output voltage will produce a low-voltage level as indicated by the arrows, identified with the reference numerals 30 and 32. In this manner, the data rate at which the signal to noise ratio (S/N) of the photoreceiver 20 is adequate will be dependent on the ability of the transimpedance amplifier 26 to push and pull current in and out of its input through the photodiodes 22 and 24, which, in turn, is dependent on the magnitude of photocurrent induced in the photodetectors 22 and 24. Because of the high shot noise produced by the photodiodes 22 and 24, the photoreceiver 20 normally requires a large amount of optical gain in front of the photodiodes 22, 24 and a correspondingly high optical power in order to reduce the noise contributions of the photodiodes 22 and 24 as well as the transimpedance amplifier 26. Unfortunately, known transimpedance amplifier circuit topologies have common emitter or common-source inputs and cannot equally source and sink current at the same rate without incurring significant degregation to the input waveform, especially at high optical power levels which results in degregation of the bit error rate of the photoreceiver 20. Thus, in order to achieve a relatively high data rate, the photodiodes 22, 24 are AC coupled to a 50 .OMEGA. shunt resistor, connected between a common node 34 between the photodiodes 22, 24 and ground, for providing a means for sourcing and sinking the optical induced photocurrents of the photodiodes 22 and 24.
There are several disadvantages of the photoreceiver 20 illustrated in FIG. 1. For example, the 50 .OMEGA. DC shunt resistor 36 used to sink and source the modulated photocurrents induced by the photodiodes 22 and 24, adds substantial noise, for example, 18 pA/sqrt (Hz). In addition, the AC coupling capacitor 28 inhibits DC biased modulation of the transimpedance amplifier 26 which causes the low frequency response of the photoreceiver 20 to be limited by the size of the AC coupling capacitor used. In many known systems, the AC coupling capacitor is substantially large in value which consequently prevents the entire photoreceiver 20 from being monolithically integrated in a self-contained microwave monolithic integrated circuit (MMIC). In order to eliminate such problem, direct coupled DPSK receivers have been developed. An example of such a direct coupled DPSK receiver is described and illustrated in "High Sensitivity Optically Pre-Amplified Direct Detection DPSK Receiver With Active Delay Line Stabilization", supra and illustrated in FIGS. 2a.
The direct coupled photoreceiver illustrated in FIG. 2a, generally identified with the reference numeral 40, includes a pair of balanced photodiodes 42, 44 and a transimpedance amplifier 46, configured for push pull operation. FIGS. 2b and 2c illustrate the push/pull operation of the receiver 40 in response to light excitation as indicated by the arrows 48 and 50. A schematic representation of the transimpedance amplifier 46 is illustrated in FIG. 3 and described in detail in U.S. Pat. No. 5,398,004, assigned to the same assignee as the assignee of the present invention and hereby incorporated by reference.
As shown in FIG. 3, the transimpedance amplifier 46 includes a common emitter configured input transistor Q.sub.1. For such a common emitter input transistor, an equivalent input voltage of 18 mV causes the output of the common emitter transistor Q.sub.1 to be compressed in gain by 1 dB, set by the inherent physics of the transistor which means that the input of the common emitter transistor cannot handle much current or voltage swing. Beyond the linear input power, voltage and current ranges, the transimpedance amplifier 46 has unequal charge and discharge characteristics which results in asymmetric rise and fall times of the output waveform at high speeds and thus overall effective bandwidth degregation, limited by the slower of the charging mechanisms and evident degregation of the BER (system bit error rate) of the received data stream, especially for higher incident power levels, as illustrated in FIGS. 4a, 4b, 5a, 5b, and 5c.
FIG. 4a and 4b represent graphical illustrations of the wideband output power and input impedance response of the common emitter configured input transimpedance amplifier 46 over an increasingly stepped input power level delivered from a 50 .OMEGA. source. As shown in FIG. 4a, the output power of the transimpedance amplifier 46 is nearly saturated, even at a relatively low input power -20 dBm; unacceptable for many known optically preamplified DPSK direct detection receiver applications which normally operate between -15 dBm and +5 dBm of incident optical power. FIG. 4a also reflects poor gain or output power flatness that is heavily dependant on input power level. For example, at -20 dBm input power, the output response is flat at low frequency and then begins a well-behaved roll-off response, typical of a small signal amplifier. However, at an input power level of -5 dBm, the output power starts out actually lower that the -20 dBm response and peaks up above 10 GHz where the output power has a more gradual roll off with frequency, indicative of the response of a non-linear driven amplifier. FIG. 4b illustrates that as input power is increased, the input impedance dramatically increases. Such a characteristic presents a detrimental effect of the bandwidth of the photoreceiver, whose source impedance is normally very large and can be modeled by a small shunt capacitor which makes a dominant pole with the input impedance of the transimpedance amplifier 46.
FIGS. 5a, 5b and 5c illustrate the signal distortion as a function of increased power levels. In particular, FIGS. 5a, 5b and 5c illustrate the transient response of the common emitter configured transimpedance amplifier 46 to a 10 GHz current pulse generator for increasing pulse amplitudes of 0.2 milliamps (mA), 2.0 mA and 4.0 mA, respectively. At a low current input pulse amplitude of 0.2 mA as illustrated in FIG. 5a, the input voltage maintains its square wave-shaped form and the output maintains a symmetrical sine-wave shape. As the current pulse amplitude is increased to 2.0 mA and 4.0 mA, as illustrated in FIGS. 5b and 5c, respectively, the input voltage waveform begins to distort. In particular, the input voltage waveform demonstrates an RC slewing characteristic on the falling edge when the base of the input common emitter transistor is discharging. However, there is no evidence of an RC slewing characteristic on the rising edge, which creates an asymmetry in the rise and fall waveform characteristics due to the different charging and discharge characteristic of the base, consistent with conventional bipolar theory. The slow discharging of the base to bring the transistor out of saturation is slow due to the removal of minority carrier base charge storage which must recombine in a base bulk before the voltage can follow the input waveform. The corresponding output voltage waveform also exhibits asymmetric rise and fall times as dwell times, which would propagate into jitter noise. The characteristics discussed above are typical characteristics of an overdriven amplifier, an inherent problem for conventional common emitter transimpedance amplifier topologies.
The RC slewing characteristics of the input voltage can be suppressed using a 50 .OMEGA. input resistor, but at the expense of added thermal noise. However, the output waveform is still limited in frequency response at high input levels. In particular, FIG. 5d illustrates the transient response of a common emitter transimpedance amplifier which utilizes a 50 .OMEGA. shunt resistor to accommodate the large push pull current and is also AC coupled to the photodiodes. As shown in FIG. 5d, although the input waveform exhibits a well-defined square wave, the output waveform still shows unequal rise and fall times as well as general asymmetry, which limits the amplifiers latter stages, which cannot take the large signal levels amplified by the high gain common emitter input stage. As such, high gain common emitter input stage must be followed by an amplifier which can accept a large signal without being driven into saturation.